In the present signal modulation technique represented by mobile telephones, the method to realize accurate 90.degree. phase generators in much higher frequency and much broader bandwidth became an important matter. In general, modulation arrangements for narrow band telecommunication systems using modulation, such as single sideband modulation (SSB), phase shift keying modulation (PSK), quadrature amplitude modulation (QAM), amplitude modulation phase modulation (AMPM), and a kind of frequency shift keying modulation (FSK), need two carrier signals having 90.degree. phase difference.
In such arrangements, a signal from a local oscillator is passed through a 90.degree. phase shifter to produce two quadrature phase carrier signals having 90.degree. phase difference. I-channel signal and Q-channel signal are modulated with the quadrature phase carrier signals by the mixer to produce two signals having 90.degree. phase difference. These two signals are combined at a power combiner and sent out through a band-pass-filter. At the receiver side, the received signal is shifted by 90.degree. at a 90.degree. phase shifter, and the 0.degree. received signal and the 90.degree. received signal are multiplied at a mixer to produce a received data.
There are two approaches for obtaining such two carrier signals having 90.degree. phase difference, i.e., the quadrature signal waves by an integrated circuit. One method is called RC phase shifting, which utilizes a fact that if an electric current flows a serially connected circuit of a capacitor and a resistor, then the voltage across the capacitor has 90.degree. phase difference with respect to the electric current and that the voltage across the resistor is in phase with the electric current.
The other method is called digital phase shifting, which utilizes a below fact. If we produce a signal having two times higher frequency as that of the inputted carrier signal, make it a clock signal for two T-flip flops, set one of the T-flip flops at the rising edge of the clock signal, and set the other T-flip flop at the falling edge of the clock signal, then two signal at the same frequency as the carrier signal having 90.degree. phase difference is obtained as the output signal of each of the T-flip flops.
In the 90.degree. phase generators of the prior art, a signal having two times higher frequency than the inputted frequency and 50% duty cycle is obtained from the output Z of a mixer, and two output signals having 90.degree. phase difference at the same frequency as the original frequency are obtained by cascade-connected 1/2 frequency dividers. However, it is necessary that the output Z of the mixers have sufficient bandwidth for the two times higher frequency than the original frequency and that the 1/2 frequency dividers operate accurately at the frequency.
Accordingly, input frequency which is not higher than a half of the maximum operating frequency of the circuit can be used as the maximum operating frequency for the 90.degree. phase generator. This is a drawback in the viewpoint of the current needs for high speed (FIG. 2, ISSCC98/Session 23/Paper SP23.1 p. 365).
If an ideal mixer exists and the duty cycle of its output Z can be 50%, the phase difference of its inputs X and Y would be 90.degree.. Therefore it would be possible to make a 90.degree. phase generator without 1/2 frequency dividers. However, in actual circuits, the phase errors in the input X and Y of the mixer do not have the necessary accuracy. Therefore, the circuit shown in FIG. 2 has been used in actual.
We will explain the invention using mathematical equations. If an ideal mixer was used (assuming that it exists), then its operation is written as follows. EQU Z=X.multidot.Y,
where, X and Y are the inputs, and Z is the output.
X and Y have the magnitude of 1 and the same frequency, but a different phase. Following expression can be used for X and Y. EQU X=cos(.omega.t) EQU Y=cos(.omega.t+.phi.),
where, .omega. is the angular frequency, t is the time, and .phi. is the phase difference between X and Y. The output is: ##EQU1##
The dc component Z.sub.dc is: ##EQU2##
If .phi.=90.degree., Z.sub.dc is 0, and the integrated value of Z is substantially constant. Therefore, the delay of voltage controlled delay circuit becomes constant.
However, in an actual mixer, a relative delay error (.theta..sub.e) exists between X and Y. Considering the operation of the mixer, the actual output Z' is expressed as follows. ##EQU3##
Its dc component Z'.sub.dc is: ##EQU4##
When .phi.=90.degree.-.theta..sub.e (i.e., when shifted by the amount of the phase error between the input X and Y of the mixer), as is aforementioned, the output of the voltage controlled delay circuit becomes constant, and the output of the 90.degree. phase generator will be a value including the error of .theta..sub.e.
Accordingly, in a 90.degree. phase generator using a conventional mixer, there is a problem on its operating frequency or its accuracy.
The purpose of the present invention is to eliminate the disadvantage of such a conventional circuit and easily implement a 90.degree. phase generator which can operate at higher frequency while maintaining high accuracy.